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A μA741 integrated circuit, one of the most successful operational amplifiers  
Type 
Discrete circuit Integrated circuit 

Invented  Karl D. Swartzel Jr. 
First production  1941 
Pin configuration 

Electronic symbol  
Circuit diagram symbol for an opamp. Pins are labeled as listed above. 
An operational amplifier (often opamp or opamp) is a DCcoupled highgain electronic voltage amplifier with a differential input and, usually, a singleended output.^{[1]} In this configuration, an opamp produces an output potential (relative to circuit ground) that is typically hundreds of thousands of times larger than the potential difference between its input terminals. Operational amplifiers had their origins in analog computers, where they were used to perform mathematical operations in many linear, nonlinear, and frequencydependent circuits.
The popularity of the opamp as a building block in analog circuits is due to its versatility. Due to negative feedback, the characteristics of an opamp circuit, its gain, input and output impedance, bandwidth etc. are determined by external components and have little dependence on temperature coefficients or manufacturing variations in the opamp itself.
Opamps are among the most widely used electronic devices today, being used in a vast array of consumer, industrial, and scientific devices. Many standard IC opamps cost only a few cents in moderate production volume; however, some integrated or hybrid operational amplifiers with special performance specifications may cost over US$100 in small quantities.^{[2]} Opamps may be packaged as components or used as elements of more complex integrated circuits.
The opamp is one type of differential amplifier. Other types of differential amplifier include the fully differential amplifier (similar to the opamp, but with two outputs), the instrumentation amplifier (usually built from three opamps), the isolation amplifier (similar to the instrumentation amplifier, but with tolerance to commonmode voltages that would destroy an ordinary opamp), and negativefeedback amplifier (usually built from one or more opamps and a resistive feedback network).
The amplifier's differential inputs consist of a noninverting input (+) with voltage V_{+} and an inverting input (–) with voltage V_{−}; ideally the opamp amplifies only the difference in voltage between the two, which is called the differential input voltage. The output voltage of the opamp V_{out} is given by the equation
where A_{OL} is the openloop gain of the amplifier (the term "openloop" refers to the absence of a feedback loop from the output to the input).
The magnitude of A_{OL} is typically very large (100,000 or more for integrated circuit opamps), and therefore even a quite small difference between V_{+} and V_{−} drives the amplifier output nearly to the supply voltage. Situations in which the output voltage is equal to or greater than the supply voltage are referred to as saturation of the amplifier. The magnitude of A_{OL} is not well controlled by the manufacturing process, and so it is impractical to use an openloop amplifier as a standalone differential amplifier.
Without negative feedback, and perhaps with positive feedback for regeneration, an opamp acts as a comparator. If the inverting input is held at ground (0 V) directly or by a resistor R_{g}, and the input voltage V_{in} applied to the noninverting input is positive, the output will be maximum positive; if V_{in} is negative, the output will be maximum negative. Since there is no feedback from the output to either input, this is an openloop circuit acting as a comparator.
If predictable operation is desired, negative feedback is used, by applying a portion of the output voltage to the inverting input. The closedloop feedback greatly reduces the gain of the circuit. When negative feedback is used, the circuit's overall gain and response becomes determined mostly by the feedback network, rather than by the opamp characteristics. If the feedback network is made of components with values small relative to the op amp's input impedance, the value of the opamp's openloop response A_{OL} does not seriously affect the circuit's performance. The response of the opamp circuit with its input, output, and feedback circuits to an input is characterized mathematically by a transfer function; designing an opamp circuit to have a desired transfer function is in the realm of electrical engineering. The transfer functions are important in most applications of opamps, such as in analog computers. High input impedance at the input terminals and low output impedance at the output terminal(s) are particularly useful features of an opamp.
In the noninverting amplifier on the right, the presence of negative feedback via the voltage divider R_{f}, R_{g} determines the closedloop gain A_{CL} = V_{out} / V_{in}. Equilibrium will be established when V_{out} is just sufficient to "reach around and pull" the inverting input to the same voltage as V_{in}. The voltage gain of the entire circuit is thus 1 + R_{f}/R_{g}. As a simple example, if V_{in} = 1 V and R_{f} = R_{g}, V_{out} will be 2 V, exactly the amount required to keep V_{−} at 1 V. Because of the feedback provided by the R_{f}, R_{g} network, this is a closedloop circuit.
Another way to analyze this circuit proceeds by making the following (usually valid) assumptions:^{[3]}
The input signal V_{in} appears at both (+) and (−) pins, resulting in a current i through R_{g} equal to V_{in}/R_{g}:
Since Kirchhoff's current law states that the same current must leave a node as enter it, and since the impedance into the (−) pin is near infinity, we can assume practically all of the same current i flows through R_{f}, creating an output voltage
By combining terms, we determine the closedloop gain A_{CL}:
An ideal opamp is usually considered to have the following characteristics:^{[4]}^{[5]}
The first rule only applies in the usual case where the opamp is used in a closedloop design (negative feedback, where there is a signal path of some sort feeding back from the output to the inverting input). These rules are commonly used as a good first approximation for analyzing or designing opamp circuits.^{[6]}^{:177}
None of these ideals can be perfectly realized. A real opamp may be modeled with noninfinite or nonzero parameters using equivalent resistors and capacitors in the opamp model. The designer can then include these effects into the overall performance of the final circuit. Some parameters may turn out to have negligible effect on the final design while others represent actual limitations of the final performance that must be evaluated.
Real opamps differ from the ideal model in various aspects.
Real operational amplifiers suffer from several nonideal effects:
The opamp gain calculated at DC does not apply at higher frequencies. Thus, for highspeed operation, more sophisticated considerations must be used in an opamp circuit design.
Modern integrated FET or MOSFET opamps approximate more closely the ideal opamp than bipolar ICs when it comes to input impedance and input bias currents. Bipolars are generally better when it comes to input voltage offset, and often have lower noise. Generally, at room temperature, with a fairly large signal, and limited bandwidth, FET and MOSFET opamps now offer better performance.
Sourced by many manufacturers, and in multiple similar products, an example of a bipolar transistor operational amplifier is the 741 integrated circuit designed in 1968 by David Fullagar at Fairchild Semiconductor after Bob Widlar's LM301 integrated circuit design.^{[10]} In this discussion, we use the parameters of the Hybridpi model to characterize the smallsignal, grounded emitter characteristics of a transistor. In this model, the current gain of a transistor is denoted h_{fe}, more commonly called the β.^{[11]}
A smallscale integrated circuit, the 741 opamp shares with most opamps an internal structure consisting of three gain stages:^{[12]}
Additionally, it contains current mirror (outlined red) bias circuitry and compensation capacitor (30 pF).
The input stage consists of a cascaded differential amplifier (outlined in blue) followed by a currentmirror active load. This constitutes a transconductance amplifier, turning a differential voltage signal at the bases of Q1, Q2 into a current signal into the base of Q15.
It entails two cascaded transistor pairs, satisfying conflicting requirements. The first stage consists of the matched NPN emitter follower pair Q1, Q2 that provide high input impedance. The second is the matched PNP commonbase pair Q3, Q4 that eliminates the undesirable Miller effect; it drives an active load Q7 plus matched pair Q5, Q6.
That active load is implemented as a modified Wilson current mirror; its role is to convert the (differential) input current signal to a singleended signal without the attendant 50% losses (increasing the opamp's openloop gain by 3 dB).^{[nb 4]} Thus, a smallsignal differential current in Q3 versus Q4 appears summed (doubled) at the base of Q15, the input of the voltage gain stage.
The (classA) voltage gain stage (outlined in magenta) consists of the two NPN transistors Q15/Q19 connected in a Darlington configuration and uses the output side of current mirror Q12/Q13 as its collector (dynamic) load to achieve its high voltage gain. The output sink transistor Q20 receives its base drive from the common collectors of Q15 and Q19; the levelshifter Q16 provides base drive for the output source transistor Q14.
The transistor Q22 prevents this stage from delivering excessive current to Q20 and thus limits the output sink current.
The output stage (Q14, Q20, outlined in cyan) is a Class AB complementarysymmetry amplifier. It provides an output drive with impedance of ≈50Ω, in essence, current gain. Transistor Q16 (outlined in green) provides the quiescent current for the output transistors, and Q17 provides output current limiting.
Provide appropriate quiescent current for each stage of the opamp.
The resistor (39 kΩ) connecting the (diodeconnected) Q11 and Q12, and the given supply voltage (V_{S+}−V_{S−}), determine the current in the current mirrors, (matched pairs) Q10/Q11 and Q12/Q13. The collector current of Q11, i_{11} * 39 kΩ = V_{S+} − V_{S−} − 2 V_{BE}. For the typical V_{S} = ±20 V, the standing current in Q11/Q12 (as well as in Q13) would be ≈1 mA. A supply current for a typical 741 of about 2 mA agrees with the notion that these two bias currents dominate the quiescent supply current.
Transistors Q11 and Q10 form a Widlar current mirror, with quiescent current in Q10 i_{10} such that ln( i_{11} / i_{10} ) = i_{10} * 5 kΩ / 28 mV, where 5 kΩ represents the emitter resistor of Q10, and 28 mV is V_{T}, the thermal voltage at room temperature. In this case i_{10} ≈ 20 μA.
The biasing circuit of this stage is set by a feedback loop that forces the collector currents of Q10 and Q9 to (nearly) match. The small difference in these currents provides the drive for the common base of Q3/Q4 (note that the base drive for input transistors Q1/Q2 is the input bias current and must be sourced externally). The summed quiescent currents of Q1/Q3 plus Q2/Q4 is mirrored from Q8 into Q9, where it is summed with the collector current in Q10, the result being applied to the bases of Q3/Q4.
The quiescent currents of Q1/Q3 (resp., Q2/Q4) i_{1} will thus be half of i_{10}, of order ≈ 10 μA. Input bias current for the base of Q1 (resp. Q2) will amount to i_{1} / β; typically ≈50 nA, implying a current gain h_{fe} ≈ 200 for Q1(Q2).
This feedback circuit tends to draw the common base node of Q3/Q4 to a voltage V_{com} − 2 * V_{BE}, where V_{com} is the input commonmode voltage. At the same time, the magnitude of the quiescent current is relatively insensitive to the characteristics of the components Q1–Q4, such as h_{fe}, that would otherwise cause temperature dependence or parttopart variations.
Transistor Q7 drives Q5 and Q6 into conduction until their (equal) collector currents match that of Q1/Q3 and Q2/Q4. The quiescent current in Q7 is V_{BE} / 50 kΩ, about 35μA, as is the quiescent current in Q15, with its matching operating point. Thus, the quiescent currents are pairwise matched in Q1/Q2, Q3/Q4, Q5/Q6, and Q7/Q15.
Quiescent currents in Q16 and Q19 are set by the current mirror Q12/Q13, which is running at ≈ 1 mA. Through some^{[vague]} mechanism, the collector current in Q19 tracks that standing current.
In the circuit involving Q16 (variously named rubber diode or V_{BE} multiplier), the 4.5 kΩ resistor must be conducting about 100 μA, with the Q16 V_{BE} roughly 700 mV. Then the V_{CB} must be about 0.45 V and V_{CE} at about 1.0 V. Because the Q16 collector is driven by a current source and the Q16 emitter drives into the Q19 collector current sink, the Q16 transistor establishes a voltage difference between Q14 base and Q20 base of ≈ 1 V, regardless of the commonmode voltage of Q14/Q20 base. The standing current in Q14/Q20 will be a factor exp(100 mV / V_{T} ) ≈ 36 smaller than the 1 mA quiescent current in the class A portion of the op amp. This (small) standing current in the output transistors establishes the output stage in class AB operation and reduces the crossover distortion of this stage.
A small differential input voltage signal gives rise, through multiple stages of current amplification, to a much larger voltage signal on output.
The input stage with Q1 and Q3 is similar to an emittercoupled pair (longtailed pair), with Q2 and Q4 adding some degenerating impedance. The input impedance is relatively high because of the small current through Q1Q4. A typical 741 op amp has an differential input impedance of about 2 MΩ. The common mode input impedance is even higher, as the input stage works at an essentially constant current.
A differential voltage V_{In} at the opamp inputs (pins 3 and 2, respectively) gives rise to a small differential current in the bases of Q1 and Q2 i_{In} ≈ V_{In} / ( 2 h_{ie} * h_{fe}). This differential base current causes a change in the differential collector current in each leg by i_{In} * h_{fe}. Introducing the transconductance of Q1, g_{m} = h_{fe} / h_{ie}, the (smallsignal) current at the base of Q15 (the input of the voltage gain stage) is V_{In} * g_{m} / 2.
This portion of the op amp cleverly changes a differential signal at the op amp inputs to a singleended signal at the base of Q15, and in a way that avoids wastefully discarding the signal in either leg. To see how, notice that a small negative change in voltage at the inverting input (Q2 base) drives it out of conduction, and this incremental decrease in current passes directly from Q4 collector to its emitter, resulting in a decrease in base drive for Q15. On the other hand, a small positive change in voltage at the noninverting input (Q1 base) drives this transistor into conduction, reflected in an increase in current at the collector of Q3. This current drives Q7 further into conduction, which turns on current mirror Q5/Q6. Thus, the increase in Q3 emitter current is mirrored in an increase in Q6 collector current; the increased collector currents shunts more from the collector node and results in a decrease in base drive current for Q15. Besides avoiding wasting 3 dB of gain here, this technique decreases commonmode gain and feedthrough of power supply noise.
A current signal i at Q15's base gives rise to a current in Q19 of order i * β^{2} (the product of the h_{fe} of each of Q15 and Q19, which are connected in a Darlington pair). This current signal develops a voltage at the bases of output transistors Q14/Q20 proportional to the h_{ie} of the respective transistor.
Output transistors Q14 and Q20 are each configured as an emitter follower, so no voltage gain occurs there; instead, this stage provides current gain, equal to the h_{fe} of Q14 (resp. Q20).
The output impedance is not zero, as it would be in an ideal opamp, but with negative feedback it approaches zero at low frequencies.
The net openloop smallsignal voltage gain of the op amp involves the product of the current gain h_{fe} of some 4 transistors. In practice, the voltage gain for a typical 741style op amp is of order 200,000, and the current gain, the ratio of input impedance (≈2−6 MΩ) to output impedance (≈50Ω) provides yet more (power) gain.
The ideal op amp has infinite commonmode rejection ratio, or zero commonmode gain.
In the present circuit, if the input voltages change in the same direction, the negative feedback makes Q3/Q4 base voltage follow (with 2V_{BE} below) the input voltage variations. Now the output part (Q10) of Q10Q11 current mirror keeps up the common current through Q9/Q8 constant in spite of varying voltage. Q3/Q4 collector currents, and accordingly the output current at the base of Q15, remain unchanged.
In the typical 741 op amp, the commonmode rejection ratio is 90 dB, implying an openloop commonmode voltage gain of about 6.
The innovation of the Fairchild μA741 was the introduction of frequency compensation via an onchip (monolithic) capacitor, simplifying application of the op amp by eliminating the need for external components for this function. The 30 pF capacitor stabilizes the amplifier via Miller compensation and functions in a manner similar to an opamp integrator circuit. Also known as 'dominant pole compensation' because it introduces a pole that masks (dominates) the effects of other poles into the open loop frequency response; in a 741 op amp this pole can be as low as 10 Hz (where it causes a −3 dB loss of open loop voltage gain).
This internal compensation is provided to achieve unconditional stability of the amplifier in negative feedback configurations where the feedback network is nonreactive and the closed loop gain is unity or higher. By contrast, amplifiers requiring external compensation, such as the μA748, may require external compensation or closedloop gains significantly higher than unity.
The "offset null" pins may be used to place external resistors (typically in the form of the two ends of a potentiometer, with the slider connected to V_{S–}) in parallel with the emitter resistors of Q5 and Q6, to adjust the balance of the Q5/Q6 current mirror. The potentiometer is adjusted such that the output is null (midrange) when the inputs are shorted together.
The transistors Q3, Q4 help to increase the reverse V_{BE} rating: the baseemitter junctions of the NPN transistors Q1 and Q2 break down at around 7V, but the PNP transistors Q3 and Q4 have V_{BE} breakdown voltages around 50 V.^{[13]}
Variations in the quiescent current with temperature, or between parts with the same type number, are common, so crossover distortion and quiescent current may be subject to significant variation.
The output range of the amplifier is about one volt less than the supply voltage, owing in part to V_{BE} of the output transistors Q14 and Q20.
The 25 Ω resistor at the Q14 emitter, along with Q17, acts to limit Q14 current to about 25 mA; otherwise, Q17 conducts no current.
Current limiting for Q20 is performed in the voltage gain stage: Q22 senses the voltage across Q19's emitter resistor (50Ω); as it turns on, it diminishes the drive current to Q15 base.
Later versions of this amplifier schematic may show a somewhat different method of output current limiting.
While the 741 was historically used in audio and other sensitive equipment, such use is now rare because of the improved noise performance of more modern opamps. Apart from generating noticeable hiss, 741s and other older opamps may have poor commonmode rejection ratios and so will often introduce cableborne mains hum and other commonmode interference, such as switch 'clicks', into sensitive equipment.
The "741" has come to often mean a generic opamp IC (such as μA741, LM301, 558, LM324, TBA221 — or a more modern replacement such as the TL071). The description of the 741 output stage is qualitatively similar for many other designs (that may have quite different input stages), except:
Opamps may be classified by their construction:
IC opamps may be classified in many ways, including:
The use of opamps as circuit blocks is much easier and clearer than specifying all their individual circuit elements (transistors, resistors, etc.), whether the amplifiers used are integrated or discrete circuits. In the first approximation opamps can be used as if they were ideal differential gain blocks; at a later stage limits can be placed on the acceptable range of parameters for each opamp.
Circuit design follows the same lines for all electronic circuits. A specification is drawn up governing what the circuit is required to do, with allowable limits. For example, the gain may be required to be 100 times, with a tolerance of 5% but drift of less than 1% in a specified temperature range; the input impedance not less than one megohm; etc.
A basic circuit is designed, often with the help of circuit modeling (on a computer). Specific commercially available opamps and other components are then chosen that meet the design criteria within the specified tolerances at acceptable cost. If not all criteria can be met, the specification may need to be modified.
A prototype is then built and tested; changes to meet or improve the specification, alter functionality, or reduce the cost, may be made.
That is, the opamp is being used as a voltage comparator. Note that a device designed primarily as a comparator may be better if, for instance, speed is important or a wide range of input voltages may be found, since such devices can quickly recover from full on or full off ("saturated") states.
A voltage level detector can be obtained if a reference voltage V_{ref} is applied to one of the opamp's inputs. This means that the opamp is set up as a comparator to detect a positive voltage. If the voltage to be sensed, E_{i}, is applied to op amp's (+) input, the result is a noninverting positivelevel detector: when E_{i} is above V_{ref}, V_{O} equals +V_{sat}; when E_{i} is below V_{ref}, V_{O} equals −V_{sat}. If E_{i} is applied to the inverting input, the circuit is an inverting positivelevel detector: When E_{i} is above V_{ref}, V_{O} equals −V_{sat}.
A zero voltage level detector (E_{i} = 0) can convert, for example, the output of a sinewave from a function generator into a variablefrequency square wave. If E_{i} is a sine wave, triangular wave, or wave of any other shape that is symmetrical around zero, the zerocrossing detector's output will be square. Zerocrossing detection may also be useful in triggering TRIACs at the best time to reduce mains interference and current spikes.
Another typical configuration of opamps is with positive feedback, which takes a fraction of the output signal back to the noninverting input. An important application of it is the comparator with hysteresis, the Schmitt trigger. Some circuits may use positive feedback and negative feedback around the same amplifier, for example trianglewave oscillators and active filters.
Because of the wide slew range and lack of positive feedback, the response of all the openloop level detectors described above will be relatively slow. External overall positive feedback may be applied, but (unlike internal positive feedback that may be applied within the latter stages of a purposedesigned comparator) this markedly affects the accuracy of the zerocrossing detection point. Using a generalpurpose opamp, for example, the frequency of E_{i} for the sine to square wave converter should probably be below 100 Hz.^{[citation needed]}
In a noninverting amplifier, the output voltage changes in the same direction as the input voltage.
The gain equation for the opamp is
However, in this circuit V_{−} is a function of V_{out} because of the negative feedback through the R_{1} R_{2} network. R_{1} and R_{2} form a voltage divider, and as V_{−} is a highimpedance input, it does not load it appreciably. Consequently
where
Substituting this into the gain equation, we obtain
Solving for $V_{\text{out}}$:
If $A_{\text{OL}}$ is very large, this simplifies to
The noninverting input of the operational amplifier needs a path for DC to ground; if the signal source does not supply a DC path, or if that source requires a given load impedance, then the circuit will require another resistor from the noninverting input to ground. When the operational amplifier's input bias currents are significant, then the DC source resistances driving the inputs should be balanced.^{[14]} The ideal value for the feedback resistors (to give minimal offset voltage) will be such that the two resistances in parallel roughly equal the resistance to ground at the noninverting input pin. That ideal value assumes the bias currents are well matched, which may not be true for all opamps.^{[15]}
In an inverting amplifier, the output voltage changes in an opposite direction to the input voltage.
As with the noninverting amplifier, we start with the gain equation of the opamp:
This time, V_{−} is a function of both V_{out} and V_{in} due to the voltage divider formed by R_{f} and R_{in}. Again, the opamp input does not apply an appreciable load, so
Substituting this into the gain equation and solving for $V_{\text{out}}$:
If $A_{\text{OL}}$ is very large, this simplifies to
A resistor is often inserted between the noninverting input and ground (so both inputs "see" similar resistances), reducing the input offset voltage due to different voltage drops due to bias current, and may reduce distortion in some opamps.
A DCblocking capacitor may be inserted in series with the input resistor when a frequency response down to DC is not needed and any DC voltage on the input is unwanted. That is, the capacitive component of the input impedance inserts a DC zero and a lowfrequency pole that gives the circuit a bandpass or highpass characteristic.
The potentials at the operational amplifier inputs remain virtually constant (near ground) in the inverting configuration. The constant operating potential typically results in distortion levels that are lower than those attainable with the noninverting topology.
Most single, dual and quad opamps available have a standardized pinout which permits one type to be substituted for another without wiring changes. A specific opamp may be chosen for its open loop gain, bandwidth, noise performance, input impedance, power consumption, or a compromise between any of these factors.
1941: A vacuum tube opamp. An opamp, defined as a generalpurpose, DCcoupled, high gain, inverting feedback amplifier, is first found in U.S. Patent 2,401,779 "Summing Amplifier" filed by Karl D. Swartzel Jr. of Bell Labs in 1941. This design used three vacuum tubes to achieve a gain of 90 dB and operated on voltage rails of ±350 V. It had a single inverting input rather than differential inverting and noninverting inputs, as are common in today's opamps. Throughout World War II, Swartzel's design proved its value by being liberally used in the M9 artillery director designed at Bell Labs. This artillery director worked with the SCR584 radar system to achieve extraordinary hit rates (near 90%) that would not have been possible otherwise.^{[16]}
1947: An opamp with an explicit noninverting input. In 1947, the operational amplifier was first formally defined and named in a paper^{[17]} by John R. Ragazzini of Columbia University. In this same paper a footnote mentioned an opamp design by a student that would turn out to be quite significant. This opamp, designed by Loebe Julie, was superior in a variety of ways. It had two major innovations. Its input stage used a longtailed triode pair with loads matched to reduce drift in the output and, far more importantly, it was the first opamp design to have two inputs (one inverting, the other noninverting). The differential input made a whole range of new functionality possible, but it would not be used for a long time due to the rise of the chopperstabilized amplifier.^{[16]}
1949: A chopperstabilized opamp. In 1949, Edwin A. Goldberg designed a chopperstabilized opamp.^{[18]} This setup uses a normal opamp with an additional AC amplifier that goes alongside the opamp. The chopper gets an AC signal from DC by switching between the DC voltage and ground at a fast rate (60 Hz or 400 Hz). This signal is then amplified, rectified, filtered and fed into the opamp's noninverting input. This vastly improved the gain of the opamp while significantly reducing the output drift and DC offset. Unfortunately, any design that used a chopper couldn't use their noninverting input for any other purpose. Nevertheless, the much improved characteristics of the chopperstabilized opamp made it the dominant way to use opamps. Techniques that used the noninverting input regularly would not be very popular until the 1960s when opamp ICs started to show up in the field.
1953: A commercially available opamp. In 1953, vacuum tube opamps became commercially available with the release of the model K2W from George A. Philbrick Researches, Incorporated. The designation on the devices shown, GAP/R, is an acronym for the complete company name. Two ninepin 12AX7 vacuum tubes were mounted in an octal package and had a model K2P chopper addon available that would effectively "use up" the noninverting input. This opamp was based on a descendant of Loebe Julie's 1947 design and, along with its successors, would start the widespread use of opamps in industry.
1961: A discrete IC opamp. With the birth of the transistor in 1947, and the silicon transistor in 1954, the concept of ICs became a reality. The introduction of the planar process in 1959 made transistors and ICs stable enough to be commercially useful. By 1961, solidstate, discrete opamps were being produced. These opamps were effectively small circuit boards with packages such as edge connectors. They usually had handselected resistors in order to improve things such as voltage offset and drift. The P45 (1961) had a gain of 94 dB and ran on ±15 V rails. It was intended to deal with signals in the range of ±10 V.
1961: A varactor bridge opamp. There have been many different directions taken in opamp design. Varactor bridge opamps started to be produced in the early 1960s.^{[19]}^{[20]} They were designed to have extremely small input current and are still amongst the best opamps available in terms of commonmode rejection with the ability to correctly deal with hundreds of volts at their inputs.
1962: An opamp in a potted module. By 1962, several companies were producing modular potted packages that could be plugged into printed circuit boards.^{[citation needed]} These packages were crucially important as they made the operational amplifier into a single black box which could be easily treated as a component in a larger circuit.
1963: A monolithic IC opamp. In 1963, the first monolithic IC opamp, the μA702 designed by Bob Widlar at Fairchild Semiconductor, was released. Monolithic ICs consist of a single chip as opposed to a chip and discrete parts (a discrete IC) or multiple chips bonded and connected on a circuit board (a hybrid IC). Almost all modern opamps are monolithic ICs; however, this first IC did not meet with much success. Issues such as an uneven supply voltage, low gain and a small dynamic range held off the dominance of monolithic opamps until 1965 when the μA709^{[21]} (also designed by Bob Widlar) was released.
1968: Release of the μA741. The popularity of monolithic opamps was further improved upon the release of the LM101 in 1967, which solved a variety of issues, and the subsequent release of the μA741 in 1968. The μA741 was extremely similar to the LM101 except that Fairchild's facilities allowed them to include a 30 pF compensation capacitor inside the chip instead of requiring external compensation. This simple difference has made the 741 the canonical opamp and many modern amps base their pinout on the 741s. The μA741 is still in production, and has become ubiquitous in electronics—many manufacturers produce a version of this classic chip, recognizable by part numbers containing 741. The same part is manufactured by several companies.
1970: First highspeed, lowinput current FET design. In the 1970s high speed, lowinput current designs started to be made by using FETs. These would be largely replaced by opamps made with MOSFETs in the 1980s.
1972: Single sided supply opamps being produced. A single sided supply opamp is one where the input and output voltages can be as low as the negative power supply voltage instead of needing to be at least two volts above it. The result is that it can operate in many applications with the negative supply pin on the opamp being connected to the signal ground, thus eliminating the need for a separate negative power supply.
The LM324 (released in 1972) was one such opamp that came in a quad package (four separate opamps in one package) and became an industry standard. In addition to packaging multiple opamps in a single package, the 1970s also saw the birth of opamps in hybrid packages. These opamps were generally improved versions of existing monolithic opamps. As the properties of monolithic opamps improved, the more complex hybrid ICs were quickly relegated to systems that are required to have extremely long service lives or other specialty systems.
Recent trends. Recently supply voltages in analog circuits have decreased (as they have in digital logic) and lowvoltage opamps have been introduced reflecting this. Supplies of 5 V and increasingly 3.3 V (sometimes as low as 1.8 V) are common. To maximize the signal range modern opamps commonly have railtorail output (the output signal can range from the lowest supply voltage to the highest) and sometimes railtorail inputs.
This article uses material from the Wikipedia article "Operational amplifier", which is released under the Creative Commons AttributionShareAlike License 3.0. There is a list of all authors in Wikipedia
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